+++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++ B.Ludwig@mx.Uni-Saarland.de NOTE: THE AUTHOR SHALL NOT BE HELD RESPONSIBLE IN ANY WAY FOR THE USE OF THE INFORMATIONAL CONTENT OF THIS ARTICLE, OR DAMAGES OR INJURIES ARISING FROM THE USE OF SAID CONTENT. QUAD-405 Modification (and related stuff) Vers. 11/2000 (# = major changes and additions since first posting on QUAD-WORLD, thanks to G.Hutchison for stylistic improvements!) Contents: Intro A) Input-stage 1) OP-Amp 2) Input-stage gain 3) Caps & resistors B) Output-Stage 1) Drivers 2) Dumpers C) Miscellaneous D) Power-Supply E) Summary F) Appendices I) How does the 405-circuit basically work II) The current-dumping-principle III) A note on slew-rate and bandwidth IV) The development of the 405 V) Replacement-parts --------------------------------------- Intro The 'QUAD 405' was in production from 1976 to the mid-nineties (of the last century). There were several minor changes in the circuitry during these nearly 20 years. At SN 59000 a major revision took place, and at SN 65000 a refined protection-circuit gave the opportunity to rename the amp to 'QUAD 405-2' in 1981 (see Appendix IV below). This new protection-circuit aside, all modifications (even those in later versions of the 405-2) can be applied to the early models without great expenditure. I think at least some of these modifications are in fact worth applying to the older models, just to reveal all the qualities of the 405's basic conception. And there are some other easy - and very cheap - modifications that will improve performance further (including a simple modification of the protection-circuit which will overcome the main weakness of the early 405). All the circuit-changes described below have been applied successfully to my own two 405s (SN. ~45.000, Circ. Diag. M12333, PCB 12368 iss.10) some years ago (both amps work perfectly since then) and most of them by others to theirs with similar results. Some (in sect. B below) especially aim at better performance with low-impedance loads (the 405 was designed during the blessed '8-Ohm-days', but I wanted to drive speakers with impedance-drops down to 3 Ohms). The others aim at better overall performance. Many of these (in sect. C) are just an 'upgrade' from the earliest 405 to the latest 405-2 (so they will be pointless in many cases), some are an 'anticipation' of the improved 306/520f/ 606-family circuitry [as indicated], and some are my own proposals (especially those in sect. A and B). The 405-schematics are not essential for most of the modifications since component-labels are printed onto the PCB. However: for understanding what you are doing schematics are essential. I will add some information about the circuit in passing to give an idea where further improvements might be possible - and where they are definitely not. The 'current-dumping' [CD] principle itself is no object of the following modifications (it just seems to work perfectly in the 405 and in its followers - see Appendix II below). But thanks to it, the amp is very (very!) stable and the output-stage is class-C (unbiased) and thus rather uncritical concerning modification and component-upgrade: As long as the (passive) 'CD-bridge' is balanced, particular properties of the output-stage components may vary in a considerably broad range without affecting performance. The main goal of the 405-design was: A state-of-the-art amp that is suitable for mass-production and whose properties will stay unchanged over a long period. Consequently all relevant properties of the amp are determined by design and by those properties of components only for which a sufficiently tight tolerance is guaranteed by the suppliers already (as for example the resistance of a resistor or the offset of an OP, in contrast to current-gain or saturation-voltage of a transistor). So there is no internal adjustment (quiescent-current, DC-offset etc.) necessary in the 405 and it should keep its specs over the whole lifetime (as long as the electrolytic caps do their job, of course). Don't hesitate criticising the following lines - they just sum up my thoughts and collect passages from my '405 internet correspondence' during the last years (many thanks to all correspondents!). Actually I'm not an audio-engineer, I'm just a hobbyist who enjoyed some training in physics decades ago and studied JAES, EWW, rec.audio.tech - and the QUAD-shematics (405/520/606). If anyone knows something about the further 'evolution' of current-dumping (in 606-2, 707 and 909), please let me know. Maybe we can learn something from it which applies to the 405 as well! (But as far as I know there are mainly modifications concerning the power-supply - a replacement of the original transformer by a cheaper ring-core-type f. e. - but I am not sure.) A) INPUT-STAGE (In sum: Replace OP-Amp, improve rail decoupling and reduce gain) The task of the input-stage-OP is to amplify the input-signal 15 times (+23dB, inverted), to form a ~13Hz, 12dB/oct high-pass and to adjust output-dc to zero (nothing mysterious there). - # Dont' wonder about R9 and R10 (and R11), they are only part of the voltage-limiter required for the ESL57 speakers, if you don't use these veterans you might want to short-circuit R9 to reduce influence of positive-rail rubbish a little further. Thanks to the high input-impedance of the subsequent stage R10 doesn't do any harm (so don't touch). I think the first two (and sonically most significant) steps in upgrading the 405/405-2 are: [1] to replace the veteran OP-Amps: LM301 and TL071, (sometimes even the LF351!), were hardly state of the audio-art in their time (the seventies) - and nowadays they are definitely not. And [2] to reduce gain of the input-stage. This is the ONLY way to increase the rather poor signal-to-noise ratio significantly. Further: At +23dB even more modern audio-OPs might contribute significantly to the overall distortion of the 405. Compare f. e. NE5534: ~0.01% THD (at +20dB, 1kHz, see data sheets) to the 405 (with LM301): < 0.01% (mainly second-order harmonics, measured at 1.3kHz and 10V output into 10 Ohms, see EWW, July 1978, p. 84). #As long as the OP-Amp is NOT the 301, step [2] can be applied even without step [1] (for those who believe that all OPs actually sound the same ...). 1a) Thanks to the plain, moderate-impedance-design (~10k) of that stage there is a wide choice of recent OP-amps that might fit (Fet as well as Bipolar). Most audio-relevant OPs are pin-compatible (741-style) except for dc-offset- and hf-compensation (pins 1, 8, 5): (from top) 2: inverting input 1 -+-v-+- 8 3: non-inverting input 2 -| |- 7 4: ps- 3 -| |- 6 6: output 4 -+---+- 5 7: ps+ An external compensation-capacitor was necessary for the LM301 (used in the early 405s; 3p3 from pin 1 to pin 8, for stability at gain ~10). To prevent unpredictable influence of that cap, it is safe to bend away (or even cut off) pin 1 or pin 8 (which may be internally connected for different purpose) if a new OP is inserted. Since there is no need for further dc-offset-trim in the 405 pin 5 is not connected on the PCB. What is the best replacement thus for the 301 or the 071? There is no use in looking for a low-noise OP, since thermal noise of the input-resistor (R2=22k, > 20nV/Sqrt(Hz), in series!) will be dominant in any case. Therefore even very different OPs like NE5534A (~3nV/~0.5pA), OPA604 (~10nV/~0.005pA), LM301, TL071 (~15nV) and even LF351 (~20nV) will all give about the same noise (believe it - or try it, I'll treat the noise-problem separately below!). I would strongly recommend Burr-Brown's neutral (call it 'transparent') "audio-workhorse" OPA604AP (single, not: OPA2604, double!) or the popular NE5534. To my experience both do an excellent job here. Someone told me he had thoroughly tested both (after the gain-reduction, see '2)' below) in comparison with the often recommended OP27; his results: the latter lacks some smoothness and the 604 gives a _slightly_ more prominent low-midrange/bass than the 5534 (try it - or believe it!). Of course, there are many other suitable up-to-date OPs (from Analog-devices, Linear Technologies, Texas-Instruments, National Semiconductors, Harris, Motorola, etc.), and even expensive devices like BB'S OPA627 or AD's 797 (which are widey regarded as the most neutral OPs available) could be considered (but they might need additional care to prevent them from oscillation, so it is NOT a matter of course that they will sound better than the 5534!). The choice is up to you. But: Don't expect any miracles, and be honest to yourself when evaluating different OPs - this will save time and money! The introduction of the 5534 in the late seventies was the giant step forward in audio-OP development. In an analog, low-gain, moderate-impedance audio-application (note the three reservations!) like in the 405 there is barely anything left to be improved by a different OP or even by discrete circuitry ever since (but it is still easy to make any OP sound badly by poor circuit-design - see 1b below). Most of the music we listen to nowadays has passed through more than one 5534 in the studio-equipment anyway. - Take care for correct orientation when inserting the new OP - otherwise Tr10 might smoke after a few seconds (I know what I'm talking about ...). 1b) Add two ~100nF-caps (here cheap ceramics are first choice!) from the OP's power-supply pins (7 and 4) to ground (pin 3 in this case). Solder them onto the copper-side of the PCB directly under the OP-case (since every tenth of an inch counts). Otherwise the 604, the 5534 and other 'modern' Audio-OPs will have a tendency to ringing or even to rf-oscillation (despite their better PSR-ratios!). This is not directly detectable at the output of the 405 due to the 50kHz-low-pass (R12/C6) following the OP. But it shows up f. e. in an increased sensitivity to noise from the mains (switching thermostats etc.) and it is sometimes audible as a sharp 'squeak' when the amp shuts down some seconds after power is switched off (the oscillator's 'good-bye' - try it, so the effect of adding the caps might become obvious to you immediately). An oscillating 604 or 5534 is, of course, much worse even than a non-oscillating LM301 or TL071. That's the reason why merely replacing the 301/071 by a 'superior' device usually messes up everything here (thanks to C6 the tweeters will always survive it). So DO NOT replace the OPs without adding the two caps for rf-decoupling (except for testing, of course). Although there is absolutely no need for more than 100nF (even 10nF should do since OP-decoupling at audio frequencies is no real problem in the 405 thanks to the small load by the subsequent stage), increasing the values up to 10uF or more will probably do no harm (but observe power-up-behaviour: maybe the OP will start-up too late when the caps are too large). 1c) Since most modern Audio-OPs need more quiescent-current (for their improved output-stages) than the general-purpose veterans, sometimes (not always) an improper power-off-behaviour will occur after OP-replacement (this has no further influence on normal sonic-performance, of course, but it's bad 'style' nevertheless). The 604 for example needs at least 5V and 5mA for proper operation. When the rail-voltage drops below ~+-20V (= 5V + 3k3*5mA) the overall DC-control will become ineffective and the output-voltage will drop towards the negative rail-voltage immediately (because Tr2, and consequently Tr7, do shut off when the OP does: ~+1.2V at the base of Tr2 are required for zero output) until the DC-clamp-circuit cuts in to protect the speakers (this is audible as 'bump' terminated by a sharp 'plick'). The old 301/071 and the LF351 (a low supply-current version of the more popular LF356) don't show this behaviour: they need less than 4V/1.5mA for operation and R7/R8 were chosen adequately (3k3) to guarantee this even below 10V rail-voltage (which seems to be sufficient for a soft shut-down). The recent OP176 (Analog Devices) is similar to the 071 in this respect (4V/2mA), so it might work perfectly without any further modification (I didn't try actually). # There seems to be a simple trick to cure this power-off-behaviour. I didn't actually try it myself, because I hit upon it by chance only after having already applied my 'rigorous' solution (which is a little more extravagant, see below). Different people have reported different results, so I fear this 'easy way out' is not always reliable. But you might try whether it works in your 405: Just a 4k7-resistor across the negative zener D2. This will make the negative OP-supply-voltage decrease faster at power-off than the positive (and has no influence during normal operation) and thus should help to keep the base of T2 sufficiently 'positive'. Instead of quibbling over tweaking R7 and R8 suitably I decided to eliminate the power-off-problem - once and forever - by just adding a pair of MPSA42/92 (=300V; they were just at hand, a pair of MPSA06/56 [=80V] should do it as well) as voltage-regulators: 50V >----+ 50V >----+-----------+ pos : MPSA42 npn | | | neg : MPSA92 pnp |R| |R| + | | | / c this is the pin- +-----> OP +---+---| b <- layout from top | >> | : | \ e |D| |D| |C| +-----> OP | | : | +---+ GND GND Now the resistors/zeners set only the base-voltage (nevertheless the 3k3 need not to be increased since diode current only rises from ~8mA to ~10mA), while the new transistors deliver all the current. They switch on completely when the rail-voltage drops below the zener-voltage. Consequently the 604 (or any other OP which might need even more current: 5534 f. e.) will operate properly as long as the rail-voltage is ~1V above its required minimum supply-voltage. To my experience the 'shut-down' is in fact absolutely noiseless with the new regulators: the residual-hiss just fades away about 5s after power-off. (# Maybe it is already sufficient to add the regulator for the posive rail alone. I didn't try. Please let me know if it works!) - The copper-track from the common D1/R7- (and D2/R8-) soldering-pad to the OP has to be cut through and the gap to be bridged by b-e of the transistor, then c has to be connected to the opposite end of R7/R8 (that's it, no extra wires are required). # You might place the two 100nF (from 1b above) across the zeners and take additional 10nF (or just 4n7) for further decoupling after the transistors (next to the OP-package) - but this is probably exaggerated (the same applies, I think, to using voltage-regulators - like a LM317/337-pair - instead of simplee transistors here). 2) The input-sensitivity of the 405/405-2 is much too high for most domestic applications (0.5Veff for full output swing, just because Quad wanted to keep the 405 compatible to the 0.775V-standard). There is no use at all in attenuating a signal heavily by the volume-knob of a pre-amp just for driving a stage with too much gain afterwards. Reducing gain of the 405 input-stage by factor ~3 {or ~4, values in braces - for even increased gain-reduction help yourself!} to 1.5V {2.0V} for full o. s. will not only improve convenience with most pre-amps. At the same time it will reduce input-stage-noise, the effect of the preamp's noise-floor, and even OP-distortion by 10dB {12.5dB} (of course, the relevance of the last point is debatable: but in any case gain-reduction will be more efficient here than for example any further improvement of the OP-power-supply). Thanks to the mod SN-ratio will approach the (excellent) value of the 606. This improvement is extremly significant when efficient speakers are used.- I suppose Quad reworked the input-stage for the 606-family mainly because with the 405-topology it was impossible to reduce noise without reducing input-sensitivity or input-impedance at the same time. You have to add three components directly onto the PCB (copper-side) to increase local feedback of the OP as well as overall feedback (#all three new components are required since the time-constants of the two feedback-paths - and of the input path - have to be preserved, otherwise the slope of the input-high-pass will be corrupted). Afterwards the stage will work at a gain of 4.6 (+13.5dB) {3.5 (+11dB)}, that's where good, low-gain-stable Audio-OPs are nearly unbeatable today. Don't use the LM301 - or the OP37 f.e - after the gain-reduction, they are not compensated for gain < 5 (the 5534 is stable > 3). [It is possible as well to reduce gain by increasing input resistor and cap (R3/C1) suitably (by just adding a R/C combination in series with same time-constant [f. e ~.33uF and 47k] in the cable/connector outside the 405), but this will not reduce noise to the same degree since the noise of the additional resistor will compensate for the gain reduction.] There is one MKT-capacitor (C4=47nF, connected to R6=330k) close to the OP. Add [1] C4'=100nF {150nF} (same type) across it. Three resistors are connected to pin 2 (inverting input of the OP): R3=22k (leading to the input cap C1=0.68uF), R4=22k (which leads to C2=100uF) and R6=330k (on later PCB-issues R6 and C4 have changed places). Add [2] R6'=150k {100k} across R6 and [3] R4'=10k {6k8} across R4=22k (not across R3!). That's it. You should use 1%-resistors for R6' at least (to keep the channel-balance). Tolerances of R4' and C4' are not too critical since the time-constants of the 13Hz-high-pass are subject to C2's much higher tolerance anyway (but 1%-resistors and 5%-caps are not really expensive ....). BTW: Moving the speakers by one or two inches will have more influence on frequency- and phase- response at any listening-position in any living-room than 10% or even 20% deviance of these components. Two (or three) alternatives to step [3] exist. All these versions give the same frequency- (and thus phase-) response, but they are a little more difficult to apply (since you have to replace components) - so take your choice! [3a] It is possible to reduce C2 from 100uF to 33uF {22uF} instead of reducing R4 by shunting R4' (this makes sense especially when C2 is old and has to be replaced anyway). If a fitting non-polar type is available, take it; if a polar (tantalum) cap is used (like in older 405s): connect '-' to the ground, '+' to the OP. (Maximum voltage of the caps doesn't matter here, 3V is already ample.) [3bi] The 'golden-ear-way' is to replace C2 by 2*68uF/25V {2*47uF} in series (= ~34uF {~23uF}) with '-'-taps connected, and to bias their junction to -15V by a ~270k-resistor from the negative OP-supply. This will eliminate any electrolytic/tantalum bias-problems (which were present in the original circuit). Correct biasing makes nearly every standard electrolytic/tantalum superior to any non-biased 'high-grade device'. - Don't bother that it takes more than a minute then for the output-offset to drop from about +100mV to the target area < 2mV after power-on. This does no harm and is due to the huge time-constant for adjusting the C2/C2-junction to -15V: ~200s = 270k*130u*2*pi (I decided in favour of this value just to avoid even the slightest interaction of the rail-noise with the audio-signal - see additional C2'= 100n below as well). [3bii] Since non-polar 33uF-caps (for [3a]) and 68uF/25V-caps (for [3bi]) are not very current, it might be more convenient to take a non-polar 47uF or 2*100uF (that's what I actually did) and to add R4'=43k across R4 to restore the proper time-constant. - Further add C2'=100nF (propylene) across C2, even if you don't want to change the latter. For gain=3.5 in cases (3a) and (3bi) C2 has to be ~25uF (not just 22uF or 23.5uF). So C2' might be 2.2uF or 1.5uF there (instead of 100nF) to come a little closer to the 'theoretical' value; but don't be too pedantic: even C2=22uF would give just -0.5dB at 20Hz - and the tolerance of the electrolytics is more than +-2uF anyway. I recommend inserting C2' into the former C2-position and to add the other caps [and the biasing-resistor] onto the copper-side. [#3c] The platinum-ear-version: Replace C2 by 6u8-foil {4u7} (BIG!) and change R4 to 24k {27k}. The frequency-response is very close to that of the original down to 13 Hz. At ~5Hz it will change by about +2dB and at 1Hz (where we are at -42dB already !) about +8dB etc. but this will not matter at all. These two input-stage-mods (OP-replacement and gain-reduction) together give an impressive stimulation of the otherwise a bit lifeless ("behind-the-curtain-")'405-sound'. The mods in C) and D) below are all less significant sonically. 3) A note on capacitors: Only C1, C4 and C6 might influence the sound-quality directly. If C6 is a styro-type (as it is in the old 405s at least - just to insure small tolerance), this is the best you can get, so don't touch! C1 and C4 are 'only' standard MKT-types - but nevertheless no significant improvement by 'high-grade-devices' seems to be possible here: The 'distortions' generated by C1 and by C4 - if there are any at all - will work exactly into opposite directions (inverting-amp!) and thus cancel out each other _completely_ at the summing-point as long as they are of same kind and order (there is no mystery behind that, but just '1+(-1)=0'). For this reason it is most important (IF type or brand of caps do matter at all) that C1 and C4 (and the additional C4') are of similar type (I heard about a 'professional upgrader' who always replaced C1 by a 10$-device, leaving C4 as it was - that's cheating). I would recommend to leave C1 and C4 as they are. But if you cannot resist changing something: Replace _both_ by polypropylenes (you might then prefer a single 150nF for 47nF+100nF). But do NOT increase the value of C1 - unless you want some resonance-peak at about 15Hz! #The electrolytics (C2, C5 and C10) should be replaced after about ten years because they tend to dry out (as they do in every audio-gear). For C2 see "2[3]" above. Since C10 (the bootstrapper) is always adequately biased and the circuit-design is not sensitive to its specific properties - it is inside the feedback-loop anyway -, it does't need any further consideration. Just replace it by a new one after a decade. On some later boards C10 is placed rather close to R31/31 which become very hot. So it would be better to mount the cap onto the backside of the PCB, this will increase liefetime (thanks, Lars!). # 100/120Hz-hum at the output is often caused by a faulty C5 - so replace it even earlier when necessary. Since all electrolytics in the 405 are adequately biased (at least, if you apply the above "golden-ear- mod", 3bii) there is nothing to complain about them. But if you distrust electrolytics in principle, adding 1uF (foil) across C5 might make sense (so even the last drop of rf-noise from the rails into the current-source - if there is any left - will be sucked up). Concerning C8 and C11: see section C) below. #4) Further there is no use in replacing any of the resistors by 1% metal-film or so-called audiophile parts. None of the resistor-tolerances is really critical. You might change most of the values by +-10% (often even +-20%) without changing performance significantly. The only exceptions would be those resistors that are implemented 5% or 1% by the manufacturer anyway: Frequency-response at 20Hz and overall gain will change by ~+-1dB, if you change the values next to the input by ~10%, gain will change with R16/20/21 and distortion will rise about twice the percentage R38's deviates from the intended value. Even at the input metal-film resistors will not reduce noise. If special 'audiophile' resistors could have any audible influence at all (I'll leave that point open), it would be here indeed. But at least for R3 and R6 the same applies as for C1 and C4 above: if one of them does any damage to the sound, the other will compensate for it exactly (no myth, merely math.), as long they are of the same type. In all other places the characteristics of the semiconductors f. e. have a much greater and less predictable influence - and have much bigger tolerances. B1) OUTPUT-STAGE, Drivers (In sum: if your speakers are 8 Ohm, skip this whole section B) The 'upper driver' (TR7) is part of the 'class-A-stage', and it might thus be tempting to try an upgrade from the venerable RCA 40872 (~BD244D, 5MHz) to a faster device (f. e. Motorola's 30Mhz-MJE15031, MJE15033, or Toshiba's 2SA1930). But there is absolutely no use in this kind of 'update': At very low levels (and that is: everywhere outside the audio-range) the 'pre-drivers' (Tr3/4) alone determine the maximum-speed of the stage (via C11) - and they are much faster than the driver itself. - Further it would be a VERY bad idea to upgrade the 'lower driver' (Tr8) by a faster device since it is part of the dumpers, where high speed is rather unwanted (speed is even reduced intentionally by R37/L1/[C19] or R37/L4 [later]). - So everything seems to be perfect with the 'cheap' and 'slow' drivers. B2) OUTPUT-STAGE, Dumpers The following mods are of any use only if speaker-impedance drops down to 4 Ohm or even less. Usually the 405 is not recommended for this kind of low-impedance-loads - but have a look ... This will require some 'hard work' and is a pleasure only to her (or him) who enjoys opening the toolbox (purists should skip the rest of this section to avoid heart-attack!) a) [405 only] Firstly there is a simple mod of the current-limiting circuit of the old 405 which brings it a little closer to the 405-2's characteristics (and keeps the short-circuit-protection). This will have no effect unless speaker-impedance drops below ~5 Ohm. Step 1: Replace R27/29 (15k or 8k2) by a 36V-Zener-diode (1.3W - pointing 'up' in the diagram - not 'down' like D3-D6) with 2k7 (2W) in series. So the current-limiter will still work as before at full output-swing and at short-circuit (just compare the voltage at the base of Tr5/6 at V_out = 0V and at V_out = 50V 'before' and 'after' the modification), but it will allow full ~7A with any load down to approx. 2 Ohm (output > ~14Vpeak). This maximum of 7A/35V (across the device) is still inside the SOA of up-to-date transistors as long as the signal is periodic. Of course, it is not for continuous DC, but that should be no problem since then the clamp-circuit will cut in and reduce current to the "original" short-circuit-conditions. The original limiter dropped down continuously from 7A at V_out = 50V to 3.5A at short-circuit. #See the following diagram which gives a rough picture. The power-limit into a given load is I_max * V_out at that point where the load-line crosses the limiter-characteristics (devide by 2 for P_rms, for example org. into 2 Ohm: ~4*8/2= 16W; mod_1 into 2 Ohm: 7*14/2 = 50W; org. and mod_1 into 4 Ohm: 7*36/2 = 100W). You might better print it out and draw the lines suitably to get an idea of the matter. But: The diagram is just qualitative, it is NOT intended to give the exact figures! I_max (peak) ^ + <-- 2-Ohm-load 10 | . . +. . . . . . | . + + <-- 4-0hm-load | . + + o 7 | . *+ * * * *+ o * * * | . * + o + |. * + o + | * o+ + o : 405 (sn > 29.000) 3.5|o* + + * : Mod. step 1 | + + . : Mod. step 2 (see 'c' below) | + + | ++ +----------------+----------------+----------+-----> | 15V 35V 50V V_out (peak) #It is more elegant, of course, to add the zener and to replace R24 and R26 by 120R and 420R while leaving R27 at 15k, so the current through the network is not increased, and accordingly there is no need for 2W-resistors - but this is a little more difficult to apply. In any case, there probably will not be THD < 0.01% at all 'unprotected' levels after this mod, for sure, and not 50W _continuous_ sine-drive into 2 Ohm (so you have to take some care not to overheat the amp when driving low-impedance-loads with continuous signals on the workbench!) - but there will be no interference by the protection-circuit during for example 7A-peaks into 2 Ohm (~50W). Further: 330nF (or 680nF) connected from base to emitter of TR5 and TR6 should eliminate any problems (if there are - as some people suppose) of the protection-circuit due to short pulses caused by signal peaks into highly inductive or capacitive loads - or even D3 and D4's 'switching' on and off. Maybe you want to add them to be on the safe side. b) [405/405-2] To get more current with less distortion, you can upgrade each single output-transistor (17556, 2SD424 - or even the veteran BDY77) by a pair (yes: a pair!) of up-to-date-devices. Doubling the devices will give much safer and better performance with low-impedance loads because each device will work at half the current (where their current-gain is higher) and resistive losses are reduced as well. Thanks to the uncritical class-C design of the dumper-stage (no quiescent current) this upgrade is no problem electrically (as the 606-family shows). Mechanically it has become rather easy thanks to the new TO-3P(L)/TO-264 'plastic'-packages for power-semiconductors. # A state-of-the-art choice for upgrading might be Toshiba's recent 2SC5200 (or 2SC5359) which replaced the recommended 2SC3281 in ~1997 (Motorola's improved[?] copy, MJL3281A, is still in production). They all have nearly constant dc-current-gain of about 100 from 10mA up to ~7A (with the older types gain drops from about 50 at 3A to less than 30). But unfortunately they are very fast (CGBP ~30Mhz) and thus not wholly uncritical. There is no benefit from increased dumper-speed here, on the contrary: if the dumpers open too fast, the class-A stage may be too slow (due to C8) to react in time. Test for overshot with 1kHz square-wave. Usually ~1nF (ceramic) from collector to base of Tr10 (like C19 in some issues of the 405) will help already. (If you are lucky, C19 and R41/L3 are present on your board [sn 9000 to 59000], this will put you onto the safe-side anyway). Motorola's MJL21194 and 21196 (a kind of improved 15024) are more conservative alternatives: They are not that fast (CGBP ~7MHz) and they show nice current-gain characteristics up to 5A as well (which is, obviously, more than ample, at least with double-output-devices). 'MJ' indicates TO-3 at Motorola, 'MJL' is TO-3P(L)/TO-264, so take care for the 'L' here, since all are available in 'classical' TO-3 as well. And if you don't want to double the output-devices: these TO-3-versions will be good for a slight improvement by drop-in replacement. The new TO-264s obviously don't fit into the old places, but there is an easy way to mount a pair of them: Remove the two old TO-3-devices and place the first two of the four new TO-264 next to the PCB-borders with the pins pointing right and left into opposite direction. Drill two additional 4mm-holes for the screws (there is enough room in this area of the PCB). View from Top: -------------- XX = original alloy-profile | | E C B // | | | | //XX=====|===|===|==+=========+=============================== //XX +---------+ | o | | //XX | Tr9 | |40872| | //XX | 3281 | +--|---|--+ /=XX=$ | | | | | //XX | O | | B E //XX +---------+ | //XX | //XX O | PCB //XX | //XX +---------+ | //XX | O | | E B /=XX=$ | Tr10 | | | | //XX | 3281 | +--|---|--+ //XX | | |40872| | //XX +---------+ | o | | //XX=====|===|===|==+=========+================================ // | | | B C E The second devices can conveniently be mounted on top of the first. To insure proper cooling, add a small aluminium-profile (2cm+3cm, ~8cm long, 3mm or 4mm; with 4 suitable 4.5mm-holes for the screws): Original, single TO-3: ---------------------- //XX /=XX=$ <-- screw_1 (M4*14) //XX //XX //XX __TTTTTTT__ <-- 17556 //XXXXXXXXXXXXXXX TTTTTT <-- 40872 //XXXXXXXXXXXXXXXXXXXXXXXXX //XX==^========^=============================== <-- PCB //XX | | //XX screw_2 (2 x M4*20, for T) /=XX=$ <-- screw_3 (M4*14) //XX // Modification, double TO-264: -------------------- A = additional aluminium-profile: 2|___ // | 3 //XX A /=XX=A=$ <--screw_1 (now: M4*17) //XX A T2T2T2T2 <-- second 3281 //XX AAAAAAAAAAAAAA //XX T1T1T1T1 <-- first 3281 //XXXXXXXXXXXXXXXXX TTTTTT <-- 40872 //XXXXXXXXXXXXXXXXXXXXXXXXXX //XX==========^================================== <-- PCB //XX | //XX screw_2 (1 x M3*30, for T1 and T2) /=XX=$ <-- screw_3 (M4*14, as before) //XX // Mount the 3281s close to the 40872-side to leave enough room for screw_1. Place the lower two (T1) first, then add the profile with screw_1, then the upper two (T2) and at last screw_2 (the nut on top of T2). All transistor-packages must be insulated with suitable pads and heat-transfer-compound has to be used (even between the profile and the top of T1). To my experience the cooling of all four devices is excellent then. Connect collector and base from T2 to those of T1 and then both by short links (across the border of the PCB) to the copper-pads of the PCB (a common 10R/2W base-resistor is recommended, see C5) below). Connect each emitter by 0.1R/2W for reliable current-sharing (so it will not be necessary to match the pairs precisely). Use insulation tubes on all links! c) Current-limiter, Step 2: Thanks to the output-pairs you can allow a maximum current of ~10A (=100W into 2 Ohm and 200W into 4 Ohm) and a short-circuit-current of ~5A (I don't think that further increase of the maximum current makes any good sense). In the original 405 reduce the 0.091R (.08R, < sn. 29000) current-sensing resistors (R35/36) to about 0.06R by soldering 0.22R/1W (0.27R/1W) across. In the 405-2 two 0.82R/1W resistors across R35 and R36 will reduce them from 0.18R to ~0.15R, this will allow ~10A instead of ~8A. The 405 will stay proteced against all kinds of electrical short-time-stress after these mods. But long-time overload or even short-circuit (with input-signal applied) will kill it thermically (that was, of course, not different with the original limiters). Since this will take some time it can usually be avoided by the careful user. # And if you don't like current-limiters at all (but why not?): just short-circuit R35 and R36 [405-2] or e-b of Tr5 and Tr6 [405]. Then the limiters are absolutely out of operation and hence there is no further improvement possible by removing any of the components (as long as the current-limiters themselves are not broken, of course)! You may thus add a simple current-limiter-switch onto the PCB temporarily if you want to try the audible influence (but keep all wires as short as possible!). Since second-breakdown limit of a pair of 3281s at 50V is < 8A, a simple voltage-independent 10A-protection is not sufficient for short-term short-circuit-safety; so removing R27/28 [in the 405] is not very useful. I admit, this double-3281-output-stage looks more like a re-design ('505'?) than a mere modification of the 405. Due to the high current-gain of the 3281s (and their successors) it might nearly match the current-capabilities of the 606's triple-17556 output-stage now. Since the driver-stages of both amps are almost identical (see C1 below) and the 405 power-supply is not too poor (2*10.000uF [405] instead of 4*6.800uF [606]), the low-impedance-performance of the modified 405 should come rather close to that of the 606 (but observe C5 below!). # The 520f (a precursor of the 606) has a double-17556 output-stage (and 4*6.800uF PS). C) MISCELLANEOUS (In Sum: One diode and a resistor should be added to the 405-I, see C5 & C6) (Here are Quad's own updates in later 405/405-2 versions; maybe some of these improvements are far below audibility, but why not be on the safe side? - see Appendix IV below) 1) [405] If R23 is 1k2 (not 3k3 as it was in SN < ~1500) then C11 should be ~1nF (not just 330p); so add 680p across C11 (the smaller value dates from early versions where each of the low-voltage 'pre-drivers' had its own collector-resistor and -cap). It is also possible (but not necessary!) to 'upgrade' to the straightforward '606-solution': Replace the two ZTX504 (Tr3/4) by high-voltage-types MPSA93 (or 92 - observe EBC pin-layout!) and connect both collectors directly to that of the 40872 (that is, short-circuit C11 and remove R23: their only task was to keep off the high-voltage output-swing from the low-voltage pre-drivers and to couple all the collectors in the ultrasonic range [for speed and stability] at the same time). 2) [405] Before sn. 59000 the feedback-capacitor (C8) was connected to the collector of the current-source (T1). The more adequate point (with respect to bridge-balance) is the collector of the input- transistor (T2). Just connect C8 to to the opposite end of R17/C7 (that is: to the end more close to the output-device-side of the PCB): One 'cut' next to C8 and one new link with ~1cm of insulated wire. I do not think that replacement of C8 by another type will make sense. [NB: There was a huge power-on-bump and a soft power-off-'crackling' in one channel of a 405 I bought second-hand some time ago. The cause was a faulty C8 which measured ok, but obviously allowed some small DC-break-through which was sufficient to open Tr3 as long as the current-source Tr1 was cut off at start-up and shutdown. After replacing C8 everything was ok!] 3) [405] In these pre-59000-versions C5, R14, R15, R18 and R22 were connected to the emitter of TR5. But they all should be on the same potential as the emitter of TR7 (otherwise there is some unwanted feedback via R35 - a serious layout-flaw, also to be found in the 520f and in early 606 issues!). - According to the schematics it looks as if you only had to cut the link between R22 and TR5 (emitter) and connect R22 to TR7 (emitter). # Unfortunately things are a little more complicated in real life. But now there is a very elegant solution nevertheless (it was submitted by P.Nunes SP, Brazil): 1) Just lift off a) the emitter of TR5 (i.e, remove TR5 and resolder only the base and collector of it, leaving the emitter out of the appropriate whole), b) the corresponding end of R35 and c) the positive supply wire (red). 2) Solder them together on a small metal strap (on the component side, of course!). 3) Link the former position of the red wire to the opposite end of R35 with ~1cm of insulated wire (on the copper side). That's it: The job is done in a straightforward way without having to cut any track. Thanks! # The only drawback of this simple solution is that now R7 has moved to the wrong end of R35. So it will sense about 0.3V "signal-induced- ripple" in addition at full current-output. But I think the zener (together with the new caps and the OP's power-supply-rejection of about 100dB inside the audio-range) will compensate for that easily. If you disaggree, you might want to change this as well. 4) [405] Later, two diodes (1N4003) were added across e-c of the output-transistors to protect them against reverse-voltage due to clipping with inductive loads. They are not intended to have any influence during normal operation. Add them if you want to drive the 405 to its output-voltage-limits - or if you want to be on the safe side anyhow. 5) IMPORTANT! [405/405-2] To improve the voltage-transfer- characteristics of the unbiased class-C dumpers (i. e. to unburden the class-A-stage) Quad in later versions of the 405-2 (and in the 520f/ 606-family) added one diode (=+0.6V) between the bases of Tr8 ad Tr9 (they called it D13; in the first 405-2 D13 was not yet introduced, but the base of Tr9 was connected to the opposite end of D6 - which was a first step into the same direection). This further reduces distortion especially at low levels and high-frequencies and should ABSOLUTELY be applied to every 405 (see Appendix F1 below). Due to the PCB-layout it is not straightforward to add the diode to the older 405-PCBs, but it is not very difficult either: [1] cut the short PCB-connection from D5 to D6 and bridge the gap by an additional diode (1N4003 f. e.); the 3 diodes are in series now, same 'direction'; the base of Tr9 is still connected to D6 [cathode] only; [2] cut the PCB-track from Tr9's base to D6 (best: about 5mm from the soldering point for the base of Tr9); [3] now connect the base of Tr9 by a short insulated wire (or, even better in terms of stability and bridge-balance at crossover, by a 10R/2W-resistor - like in the 606) directly to the opposite end of D6 (anode, i. e. between D6 and the new D13). That's it: D6 separates the bases of Tr8 and TR9 now and D13 is in the former position of D6. You might leave out step [1] if you feel uncomfortable with too much 'cutting and bridging', then you will get the first 405-2-version. In this early 405-2 (PCB 12565.6) the base of TR9 is already connected to the adequate point, so adding the third diode is not that important here. Since I am not acquainted with the 12565.6-PCB-layout, I cannot decide whether the expenditure will be reasonable. In any case: It doesn't matter at which end of D5 it is inserted, but the base of Tr9 has to stay connected to D6 - otherwise the amp will probably die (due to thermal runaway) when warming up sufficiently! 6) [405/405-2] In even later versions of the 405-2 a 75R resistor was added across the output-inductor (L2). This was - probably - for compensating some unwanted increase of inductance at higher frequencies (caused by Eddy-currents in the small coil; see JAES Jan. 1980, p. 12). It should be added because it is supposed to improve the rf-balance of the feedback-bridge. D) LAST NOT LEAST (power-supply) (In sum: just a little more rf-decoupling) A 330nF/400V cap across the mains will reduce influence of noisy power-lines. 470nF/150V from each of the transformer's outputs to ground (yellow to green) helps further against noise from transformer and rectifier [from 606]. The 405 should be absolutely insensitive to thermostats and other noisy devices then. The wires from the secondary-windings of the transformer to central-ground (green) and to the rectifier (yellow) can be led directly from top of the transformer to rectifier and caps. So they will be as short as possible and more remote from the signal-wires which minimises influence by radiation (this is the only reason for any mod here; it is, of course, useless to replace these short wires by "better" ones since resistive losses inside the transformer are dominant by far). The original 'cable-tree' was preferable from the production-process perspective only. The 10.000uF/63V power-supply caps should be replaced after more than 10 years of use because they become noisy by the time. This noise is not directly audible but it makes distortion increase. Electrolytics in power-supplies have a lifetime of less than 10.000 hrs. use when exposed to higher temperatures (as in the unventilated 405-case). Since electrolytics have become better, cheaper and MUCH smaller since 1975 (there is real progress in this area!) some 15.000uF/63V-devices will fit now (so the 405 will have the 606's power-supply-capacity). #Maybe even 22.000uF are possible, but I am not sure whether every rectifier will survive the much higher startup-current. Some people uprated to 22mF with success, but there is a small risk in that mod nevertheless! So you should not try it before you know some suitable replacement for the rectifier which fits into the case (this might be a problem - I didn't look for a replacement yet). Decoupling the power-supply-rails on each PC-board not only with 100nF but with 100uF will reduce resonance-effects due to the inductance of the wires from the PS as well as mutual interference between the channels via PS (class-B and class-C output-stages send much switching noise 'back' into the rails). Add 100uF (or 220uF)/63V across C15 and C16, this will be more effective than rewiring the whole PS! [from 606] You should check PCB-layout: On iss. 12368.10-boards the grounding of C15/16 is separated from the signal-groundings on the PCB. If this should not be the case with other issues, I would recommend to replace the PCB-ground-connection of C15/16 by an extra wire to the central ground-connection (at the screw next to the output-devices). Otherwise the signal-ground will be polluted by the output-stage rubbish. E) TO SUM UP: Simple replacements or add-ons: Op-amp: IC1 TL071 replace by NE5534, OPA604 or .... - or LM301 and add supply-decoupling --> (+) Resistors: R4 22k add 10k {or 6k8} across to reduce gain (*) R6 330k add 150k {or 100k} across to reduce gain (*) 1x 75 across L2 (3uH) 1x 4k7 across D2 [in case of shut-down-noise with new OP] Caps: C2 100uF [see details in the section A2) about gain-reduction] 1x - add 100nF polypropylene across C2 C4 .047uF add 100nF {or 150nF} MKT across to reduce gain (*) C5 100uF replace after ~10 years or in case of hum C10 47uF replace after ~10 years C11 330pF [if not already 1000pF add 680pF styro across] C13/14 10000uF replace after ~10 years of use C15/16 100nF add 100uF/63V across (observe polarity!) 2x - add 100nF to OP supply pins (+) 2x - add 470nF/150V from transformer outputs to ground 1x - add 330nF/400V across mains All these mods (and even those of section B above) can be applied without removing the PCBs. You only have to remove top, bottom and sides of the case (which is very easily done) to reach the relevant locations. (Take care: High voltage circuitry!!!). If you have the new components at hand, a complete 'sonical update' even of a first-generation 405 will not take much more than an afternoon, even if you do it step by step with some testing after each step (don't leave out that testing, because otherwise too much unsoldering might be necessary for fault-detecting - and this will damage the PCBs). With the exception of the PS-electrolytics (>> 10$ each) everything is VERY cheap: Resistors and caps don't count since there is absolutely no need for any precious or exotic components (that's the offspring of a sound circuit-design - and thus QUAD never cared much about components, except for reliability), an OPA604 is about $3, a NE5534 not even $1. And if you need 'low-impedance-power': Even the 5200s or 3281s are less than $3.- each (but you'll need 8 of them, of course - and more time and skill). Last not least: You are free to replace all the connectors by 'audiophile' parts and to rewire power-supply, input- and speaker-terminals by high-grade wire. But don't forget: most of the connectors in the chain from the first microphone up to your speakers are not at all 'audiophile' (but just 'professional') and the wires inside the 405 are very short compared to the rest of the whole audio-chain: Hundreds of feet in the recording studios before the signal reaches the CD, several feet of "visible" cables and of "invisible" wires on all the PCBs and at the discrete components involved in your own equipment, at least 5-10 feet from the amp's to the speaker's terminals, plus some inductors in the cross-overs and - chiefly - about 10-20 feet (ultra small-gauge: they sum up to several Ohms!) wires of the voice-coils. I wouldn't expect anything (sonically, of course!) from rewiring far less than 1% of that chain - but it's a harmless pastime anyway (as long as your amp doesn't smoke afterwards!), and the only restriction is your personal budget (it is just the same like with a golden watch: if it will give you a more comfortable feeling, you should buy one - but don't expect that it will show an improved time). Hope that helps, good luck - and trust your ears: You will notice that an expiry date for the current-dumping principle is not yet in sight (although even QUAD dumped current-dumping by now)! # Don't get my above proposals wrong (as at least one correspondent did!): They aim at improving the 405 (and at having some fun), not at creating a perfect amp. If someone tells you that the best upgrade of a 405 is to sell it,[s]he may be right, of course! So what? Corrections, criticism, further suggestions welcome! BL --------------------------------------------- Appendix I: How does the 405-circuit basically work? (For the beginner - I hope I got everything correct myself!) The simplest way to describe the function of a transistor (PNP and well as NPN) is the following (I'll explain just the NPN-case = Tr2 in the 405, PNP works identically when all voltages are inverted): Assume that a voltage of more than, say, 2V is applied between collector and emitter. Current flows from collector to emitter only if the voltage between base and emitter (called 'Vbe') is > 0.6V, otherwise the c-e-path is cut off. When the transistor is open the current c->e is much bigger than b->e ('high current-gain'), and c->e-current changes heavily when Vbe varies only slightly around the 0.6V-limit ('high transconductance'). Often an 'inverted' point of view is helpful: Whenever the c-e path of a transistor is conducting, the voltage between its base and emitter is ~0.6V. This may be considered as the 'b-e diode': since when a diode conducts (in direction of the arrow) there is a voltage-drop of 0.6V across it as well. This voltage-drop is rather current-independent; but in case of power-devices, Vbe will rise up to 1.5V with curent (as a rule of thumb: assume about 0R1 or 0R2 in series with the emitter inside the package). To the 405 now: At first ignore L1, L2, L3, (L4), C7, C8, C11, R17, R23 ('remove' these Cs, 'short-circuit' these Ls and Rs). They all are there for rf-stabilty only. Further ignore the current-limiters (Tr5, Tr6 and those resistors/diodes connected to their bases). For the sake of simplicity of argument assume a 450R-resistor (called Rc from now on) from the base of Tr3 to the positive rail. It 'replaces' the 4mA current-source (Tr1, R13-15 and C5 have to be ignored thus; see below for their 'resusciation'). Now look at Tr2 first: Assume its emitter is a at given voltage-level. When its base-voltage Vin (the input) rises above this emitter-voltage by ~0.6V, Tr2 opens (see above) and thus draws a collector-current 'I' from Rc. Consequently the voltage V at the base of Tr3 will drop from +50V downwards (by V = Rc*I) and Tr3 will open (at Vb=50-0.6=49.4V). As long as nothing else happens, current (through Rc and through R13 now as well) will rise further until the base of Tr4 is at 49.4V - and now Tr4 opens (you will observe that the base of Tr3 is at 50-1.2V in the meantime: 0.6V voltage-drop at each b-e-diode). Since nothing else happens, current will rise even further until Tr7 opens. By now we have got: b of Tr7 at 50-0.6V, b of Tr4 at 50-1.2V, b of Tr3 at 50-1.8V - the collector-current of Tr2 will be about 4mA thus [1.8V/450R =0.004A]. When Tr7 was closed, the voltage at its collector was ~-50V (because the collector is connected by R30/31 to the negative rail - ignore the diodes for the moment); and when Tr7 is completely open, its collector is at ~+50V (because the c-e resistance is very small then - call it zero - compared to R30/31). When Tr7 opens just a little (that's what we assume now), a current runs down R30/31 and generates a voltage-drop V. When this current is ~45mA the said voltage-drop across R30/31 is V = I*R = 0.045*1k12 = ~50V, that means (-50V+50V=0V): the collector of Tr7 is next to the zero-volt-level then (which will be the case when the amp is idle - and ~45mA is the 'idle-current' thus). Lets now ignore Tr8-Tr10 (the dumpers- section). The output of the amp is fed only by R38 then. At this point negative feedback comes into play: The amp's output is connected 'back' to the emitter of Tr2 by R20/21 (and L2). What happens thus when Tr7 opens? The R30/31-current rises and so the voltage at the output of the amp - and with it the voltage at the emitter of Tr2 (via R20/21). But when this voltage rises the voltage-difference between base and emitter of Tr2 decreases. And when this difference approaches 0.6V, Tr2 tends to close. But then its collector-current reduces and (see above) Tr3, Tr4, Tr7 will reduce their current too: the output-voltage will thus stop rising. At which voltage? Due to R20/21 and R16 the voltage at the emitter of Tr2 is exactly 180/(500+180)*Vout=(1/3.77)*Vout. So: when Vout= 3.77*(Vin-0.6) the voltage between base and emitter of Tr2 will be just as big (0.6V) as to open Tr2 sufficiently to allow the current that opens Tr3...Tr7 suitably. If the base voltage of Tr2 will increase by 1V the output of the amp will rise by 3.77 Volt (and everything will be stable again at this value). If the base-voltage will decrease by 1 Volt, the output will drop by 3.77V. - This is a simple, non- inverting, single-ended small-power amp (with a voltage-gain of 3.77), and since Tr7 never shuts off during the full output-swing (from -~45 to +~45V) it operates Class-A. What does the OP-Amp do? Two things: Firstly it gives additional amplification of the input-signal: ~15 in the original, so there is 15*3.77= ~56 overall gain. #Obviously the main design-idea was to add a CD-output-stage to an OP. Since the OP is assumed to give an undistorted output-swing of about +-12Vp, the CD-stage's voltage-gain had to be about 3.5 to give a suitable output-voltage for ~100W into 8 Ohms. Since OPs are indeed excellent voltage-amplifiers (at least nowadays), it was a nice idea to reduce the voltage-gain of the CD-stage as much as possible and to leave all the remaining voltage-amplification to the op-amp. Secondly the OP is responsible for DC-feedback (R5/C2 keep the audio-signals off, so only the dc-level of the output reaches the inverting-input of the OP). It adjusts its own output (and with it the base of Tr2) to give 0V at the 405-output when no signal is present (so offset depends only upon the offset- parameters of the OP - which are excellent by themselves [<< 5mV, small drift]). Since idle current through Tr2 is ~4mA (due to the value of Rc or due to the current-source Tr1) when the output is at zero, the voltage at the emitter of Tr2 is Re*4mA with Re = 130R (= 180 || 500), so Ve = 0.53V and the OP will thus set Vb of Tr2 to 0.53+0.6=1.13V for zero-output (in earlier versions the current- source delivered ~6mA [R14 was 0k56, not 0k47],so Vb was ~1.4V). -- Just a word concerning this current source: The amp would work identically with Rc=450R in its place (as we assumed until now for the sake of easy explanation), but by the very high dynamic resistance of the current-source (~50k) open-loop gain is dramatically increased: The same change of the c-e-current in Tr2 results in a change of Vbe in Tr3 more than hundred times higher than with the 450R resistor Rc. Now the dumpers. Tr9 first (positive output-swing): It opens when its base is 0.6V above its emitter (=the output of the amp). Since there is a voltage drop of 0.6V at each of the two diodes (D5/D6), Tr9 opens not before the collector of Tr7 is 0.6 + 2*0.6 = 1.8V above the output, that is when a current of 1.8V/47 = 38mA runs down R38. From then on any further positive current will be 'dumped' by Tr9, and the small class-A-amp (Tr7) has only to supply the current through R38 (38mA), the base current of TR9 (which is < 1/30 of the speaker-current) and the idle-current through R30/31 (45mA). This is less than 300mA altogether at full output. Thanks to 'bootstrapping' by C10 there is hardly any AC-current from Tr7 into R30/31. And finally the negative swing: When the speaker-current through R38 becomes less than 12mA, the collector of Tr7 is less than 0.6V (=12mA*47) above the output and thus the base-voltage of Tr8 (PNP) is - thanks to the two diodes (1.2V) - more thhan -0.6V below the output: Tr8 opens and speaker-current will thus be just the difference of the currents fed by Tr7 and Tr8. When the latter's collector-current then rises 'above' -27mA, Tr10 opens too (27mA*22R = 0.6V) and 'dumps' any further negative current. Thanks to the two diodes the collector of Tr7 is still ~+0.6V above the output when Tr8 is open. Consequently Tr7 (the A-stage) will _always_ control the speakers. Since the dumpers Tr9 and Tr8/10 switch on and off during a voltage-swing and since there is a small gap (Vbe of Tr8+Tr9 = ~1.2V) where both of them are off, they work 'Class-C'. In 'Class-B' there is always a small current (the quiescent-/idle-current) that runs through at least one of the two devices - and this improves linearity drastically. This idle-current can be determined - to illustrate it at the 405-example - by a voltage applied between the bases of Tr9 and Tr8, the 'bias-voltage' (1.2V by two additional diodes for example, then Tr9 would open just when Tr8 closes - vice versa). The reason for using 'dirty' class-C in the 405 is that class-B requires additional design-care because Vbe of bipolars is not only device-dependent but drops with temperature from ~0.6V (at 25C) to ~0.3V next to the working-temperature limit (~200C) of the transistors. So the quiescent-current in class-B heavily depends on temperature if no special thermal-control of the bias-voltage is added. And even worse: when the amp warms up, Vbe goes down, the current rises and so the amp will warm up even further and consequently Vbe goes down further ... (that's 'thermal runaway'). One further diode (D13 - just one, not two!) was added in the 405-2 (and in the 606-family) between the bases of Tr8 and Tr9. It pushes the base of Tr9 up by 0.6V, so the voltage gap (where both Tr9 and Tr8/10 are closed and Tr7 alone has to control the output) is reduced from 1.2 to 0.6V (at room-temperature) and the current through R38 is further increased by 12mA. This makes error-cancelling much easier for the class-A stage (especially with low-impedance-speakers), and yet thermally unstable 'class-B' operation will not appear before the temperature-limit of the output-devices is passed anyway. Although I don't believe that it is either measurable or audible: The 405 'improves' in principle when it warms up, since it approaches a class-B-output-stage more and more. Consequently a 'hot' 405-2 should sound best and a 'hot' 405 (without D13) should sound just like a 'cold' 405-2. - NB. Whenever any kind of 'warming-up' improvement is audible in a modern class-B (or AB) amp, this reveals poor design since many simple means have been developed in the last 30 years to cancel all the effects due to these _slow_ changes in temperature. The only parameter-change with temperature which cannot be easily compensated for is curent-gain of the output-devices, but any serious design will be insensible to these (small) changes anyway (so every state-of-the- art amp should be perfect at low temperatures as well). Only _fast_ changes of junction temperature, caused by the dynamics of the programme- material, are a serious challenge until nowadays (because they are difficult to monitor in "real-time"). - Current-Dumping deals with both of them at the same time: Crossover-distortion is not just reduced, but cancelled by the class-A stage! --------------------------------------------- Appendix II: The Current-dumping-principle (CD). Here is just a 'thought-experiment' to get an idea of the CD-principle in the 405 (see 'Electronics and Wireless World' (EWW), June/July 1978 and 'Journal of the Audio Engineering Society', Jan. 1980 for further details): 'Remove' C8, R38 and 'short-circuit' L2. This will give a fictitious next-to-perfect conventional ultra-high-feedback (via R20|21) amplifier with ample of (call it: 'infinite') loop gain and thus extremely small (call it: 'zero') distortion (even with a crappy class-C output-stage!). But, of course, this amp is impossible in real life because it will be unstable due to limited bandwidth of - and thus to phase-shift by - the output-stage (otherwise audio-amp design would just be a child's play - the 'Current-Dumping Review' in EWW sept/oct. 1983 f. e. ends up in absurd conclusions because it completely ignores the stability-problem; see Peter Walker's reply in the December issue). So you will have to add C8 again as a compensation-cap (nearly every amp - Op-amps included - has a cap like this in its voltage-gain-stage, usually in a place that corresponds to c-b of Tr7 [the 'pole-splitting-capacitor']). Assume - just for the sake of the following illustrations - C8 were connected to the emitter of Tr2 (not to its collector [and to the base of Tr3] as it actually is). This makes no difference in principle as long as R12/C6 limit input- bandwidth sufficiently (# hence C6 is ABSOLUTELY required, otherwise the source-impedance for Tr3 would be too high at rf, and thus overshot or ringing were to be expected with square-wave-signals). C8 will reduce loop-gain of the driver-stage at high frequencies (by -6dB/oct), and consequently give a stable, but now only mediocre real-world amplifier: Overall loop-gain is to small now to reduce output-stage distortion adequately, and the two 'feedback-paths' (via R20|21 and via C8) are not matched as well, so cancelling the output-stage-distortion by feedback is impossible even in theory. Conventional engineering thus tries to improve the output-stage itself (f. e. by sophisticated AB-biasing-techniques). In 1975 Albinson/Walker invented a different (the CD-)solution: Adding R38 and inserting L2. If the 'square' formed by C8, R20|21, R38, L2 (the 'bridge') is balanced according to L2 = R20|21*R38*C8, the voltage at the emitter of Tr2 (C8,R20|21) is always strictly proportional to that at the output (R38,L2): Overall [!] feedback is absolutely perfect now at any frequency, even with the unavoidable compensation-cap C8 in its place, and stability is further improved by L2. If the output-stage tends to distort (especially at crossover), the driver (Tr7) will fill in the suitable correction-signal via R38, and thus the poor voltage-transfer characteristics of the class-C 'dumpers' has no influence on performance at all - as long as the driver is not overloaded, of course. Consequently the quality of the amp depends exclusively upon the linearity of the Tr7 class-A stage and upon bridge-balance. It is thus - in theory - possible to get a stable, zero-distortion power-amp even with a very robust and dirty class-C output. To give a slightly different picture: The class-C dumpers carry the output into the target-area of the low-power 'single-ended-class-A-stage' - and the latter 'makes the sound'. #This was indeed the way the 405 was advertised by Quad. But it is important nevertheless to note that the CD-principle is above all a means to compensate for the unavoidable "compensation-cap" (C 8): With a purely resistive "bridge" Current-Dumping would be entirely pointless. But when the voltage-gain-stage is an integrator (as it is necessarily, for stability reasons), the bridge-component between the dumpers and the speaker will be a small inductor which has no relevant effect inside the audio-range: Then CD has undisputable merits. In practise some (trial-and-error-)trimming of the bridge was necessary due to C8's move from Tr2's emitter to the collector (for further increased stability), the finite conductance of Tr7, the limited current-gain of the dumpers near crossover, the presence of R12, R30, C10 etc. etc. For these - and/or other - reasons there is a ~10% correction to be found in the 405-design: 0k5 * 0k047 * 0.12nF = 2.8uH (not 3.0uH as actually fitted). Quad used 5%-components, so residual bridge-unbalance will be less than 10%. This is not very much since distortion actually seems to be affected by about the same order (hence ~0.012% instead of 0.010% f. e.), and this was reasonably supposed - by the Acoustical Mfg. at least - to be inaudible (just don't overlook: even with L2 short-circuited [= 'infinite' unbalance!] distortion is still below 0.3% [up to 10kHz!, compared to < 0.01% at optimum balance]). - Of course, if you can measure crossover- distortion precisely (by scope f. e.), you might try to adjust C8 or R38 of each 405-channel individually for the last grain of improvement (for example: 4.7pF across C8 or alternatively 1k across R38 will affect bridge-balance by about +-5%). Do not change R20/21 since they affect the overall gain and thus stereo channel-balance as well. --------------------------------------------- Appendix III: A note on SLEW-RATE and BANDWIDTH A large C8 (and with it a large L2) increases stability - but at cost of power-bandwidth and/or slew-rate: C8 has to be charged and discharged correctly. Charging C8 from the output is no problem, since impedance is negligible. It is the positive input-side alone that sets the limit: With 4mA (405-2) from the Tr1-current-source the maximum positive loading-rate for C8 is: 4mA/0.12nF = 33V/us. Since the negative loading through Tr2 is faster it can be ignored. For a 1-volt change at the output (8 Ohm-load, R38=47, worst-case: dumpers are off) the collector of Tr7 has to change by (47+8)/8 * 1V = 6.8V. Hence the slew-rate limit of the 405-output into 8 Ohm is 33/6.8 = ~5V/us (the 405 was thus correctly advertised for ~0.1V/us max. input slew-rate). This looks meagre compared to any modern FET-power-amp, but nevertheless this is exactly what is needed for an undistorted, maximum-level 20kHz sine-wave, and this in turn is the fastest signal today's CD-players can deliver to the amp without overloading it: Every complex signal with 20kHz-bandwidth is either 'slower' than the full-output 20kHz-sine- wave, or it has higher amplitude and will thus drive the amp into clipping anyway (BTW: Imagine what any tweeter would do if fed with 20kHz/100W even or only the fraction of a second!). Since 'real' music is much 'slower' the 405 is more than fast enough for all relevant programme-material (high-end-mythology aside, of course). # Since all QUAD-Amps are bandwidth-limited by a ~10Hz high-pass and a ~50kHz low-pass (here: C2/R5, C6/R12), square-wave-performance does LOOK very strange when observed by scope. But as long as there is no overshot or ringing, these visible 'deformations' have nothing to do with (non-linear) distortion and are definitely inaudible. Maybe there are DC-coupled-amps with about 1MHz-bandwidth that sound different from the 405. If they actually do, there might be many reasons for that (maybe they just pass over some infra- and ultrasonic rubbish from the input to the speakers which makes them suffer!), but it definitely has nothing to do with those differences that appear in the square-wave-images. ------------------------------------------ Appendix IV: The development of the 405 from 1976 to ?? -- 405 -- Board 12368 iss. 5/6 [This is the first board described in my service-data] Board 12368 iss. 7 (SN > 2000 [?]) Emitters of Tr3 and Tr4 jointly connected to C11; C9, R19 omitted and R23: 3k3 -> 1k2 Subsonic-filter slightly modified (R4/R5: 10k/10k -> 22k/4k7) Board 12368 iss. 9 (SN > 9000) Clamp-circuit (= DC-speaker-protection) introduced (on separate PCB at the speaker-terminals) C19 (1nF) added between base and collector of Tr10, and R41/L3 (22R/6.9uH) added at collector of Tr9 (to reduce dumper-speed) C15/C16 (100nF) added for supply-decoupling on board Board 12368 iss. 10 (SN > 29000) (No board-layout-change, only change of 6 component's values) OP-supply-voltage increased from 12V to 15V (D1/D2) Slope of current-limiters slightly decreased to allow for ~100W output with 4 Ohm-loads (R35/36 0.08R -> 0.091R and R27/29 8k2 -> 15k) Board 12565 iss. 3 (major revision, SN > 59000) R14: 560R -> 470R (so current [Tr1] is reduced from 6mA to 4mA - why?) C11: 330p -> 1000p C8: connected to opposite end of R17/C7 R7, C5, R14; R15, R18 and R22: from emitter of Tr5 to emitter of Tr7 C19 (see SN 9000): omitted R41/L3(see SN 9000): omitted R37/L1 (22R/6.9uH between R36 and collector of Tr10): replaced by R37/L4 (15R/22uH/between R36 and emitter of Tr8) D10/D11 (reverse-voltage protection) 1N4003 added Main board incorporates clamp-circuit now Voltage-limiting-circuitry (only for use with old QUAD-ELS57) modified -- 405-2 -- Board 12565 iss. 4/5 (SN > 62500) First 405.2 board - fitted to 405's starting at SN 62500. Nameplate change to 405.2 at SN 65000. New protection-circuit introduced (semi-integrated-chips replace Tr5/6 and paraphernalia; R35/36 change from .091R to 0.18R) Base of Tr9 connected to opposite end of D6 Board 12565 iss. 6 (SN 66700) D13 added between D5 and D6 (base of Tr9 connected to D6/D13-joint) R44 (75R) added across L2 C20 (4.7nF) added across D2 Board 12565 iss. 7 (SN > 72500) Tr3 and R18 omitted, Tr4 changed to BC556B which gives all the current gain now. -- I actually don't understand why they didd that, since (at least up to the 707)this was not applied to the 606-design, where they did a different (and, I think, more obvious) modification (see C1 above). -- Here my service-data end -- ???? ----------------------------------------------- Appendix V: Replacement-parts Some of the original semiconductors are not available anymore (for the amateur at least). But replacements are no problem at all: BC214C (PNP, >30V, hfe>250, low noise) BC559C / BC560C / BC415C ZTX304 (NPN, >70V, ~150Mhz, hfe>50) MPSA06 / 2N5551 ZTX504 (PNP, >70V, ~150Mhz, hfe>50) MPSA56 / 2N5401 / [MPSA92] 40872 (Driver PNP, >100V, ~3MHz) BD244C / [TIP42C] 17556 (Power NPN, 150V, 15A) MJ15003 / [MJ21194] IS920 (fast-switching diode) 1N4148 (D3, D4 - speed matters!) 1N4003 (elsewhere) (Please add codes of suitable replacements for rectifier, thyristor and the Diac if you know them! I didn't look for them yet!) +++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++++